Object position and proximity detector

ABSTRACT

A proximity sensor system includes a sensor matrix array having a characteristic capacitance on horizontal and vertical conductors connected to sensor pads. The capacitance changes as a function of the proximity of an object or objects to the sensor matrix. The change in capacitance of each node in both the X and Y directions of the matrix due to the approach of an object is converted to a set of voltages in the X and Y directions. These voltages are processed by analog circuitry to develop electrical signals representative of the centroid of the profile of the object, i.e, its position in the X and Y dimensions. The profile of position may also be integrated to provide Z-axis (pressure) information.

RELATED APPLICATIONS

This application is a continuation of application Ser. No. 08/115,743,filed Aug. 31, 1993, now U.S. Pat. No. 5,374,787, which is acontinuation-in-part of co-pending application Ser. No. 7/895,934, filedJun. 8, 1992 and assigned to the same assignee as the present invention.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to object position sensing transducers andsystems. More particularly, the present invention relates to objectposition recognition useful in applications such as cursor movement forcomputing devices and other applications.

2. The Prior Art

Numerous devices are available or have been proposed for use as objectposition detectors for use in computer systems and other applications.The most familiar of such devices is the computer "mouse". Whileextremely popular as a position indicating device, a mouse hasmechanical parts and requires a surface upon which to roll its positionball. Furthermore, a mouse usually needs to be moved over long distancesfor reasonable resolution. Finally, a mouse requires the user to lift ahand from the keyboard to make the cursor movement, thereby upsettingthe prime purpose, which is usually typing on the computer.

Trackball devices are similar to mouse devices. A major difference,however is that, unlike a mouse device, a trackball device does notrequire a surface across which it must be rolled. Trackball devices arestill expensive, have moving parts, and require a relatively heavy touchas do the mouse devices. They are also large in size and doe not fitwell in a volume-sensitive application like a laptop computer.

There are several available touch-sense technologies which may beemployed for use as a position indicator. Resistive-membrane positionsensors are known and used in several applications. However, theygenerally suffer from poor resolution, the sensor surface is exposed tothe user and is thus subject to wear. In addition, resistive-membranetouch sensors are relatively expensive. A one-surface approach requiresa user to be grounded to the sensor for reliable operation. This cannotbe guaranteed in portable computers. An example of a one-surfaceapproach is the UnMouse product by MicroTouch, of Wilmington, Ma. Atwo-surface approach has poorer resolution and potentially will wear outvery quickly in time.

Resistive tablets are taught by U.S. Pat. No. 4,680,430 to Yoshikawa,U.S. Pat. No. 3,497,617 to Ellis and many others. The drawback of allsuch approaches is the high power consumption and the high cost of theresistive membrane employed.

Surface Acoustic Wave (SAW) devices have potential use as positionindicators. However, this sensor technology is expensive and is notsensitive to light touch. In addition, SAW devices are sensitive toresidue buildup on the touch surfaces and generally have poorresolution.

Strain gauge or pressure plate approaches are an interesting positionsensing technology, but suffer from several drawbacks. This approach mayemploy piezo-electric transducers. One drawback is that the piezophenomena is an AC phenomena and may be sensitive to the user's rate ofmovement. In addition, strain gauge or pressure plate approaches are asomewhat expensive because special sensors are required.

Optical approaches are also possible but are somewhat limited forseveral reasons. All would require light generation which will requireexternal components and increase cost and power drain. For example, a"finger-breaking" infra-red matrix position detector consumes high powerand suffers from relatively poor resolution.

There have been numerous attempts to provide a device for sensing theposition of thumb or other finger for use as a pointing device toreplace a mouse or trackball. Desirable attributes of such a device arelow power, low profile, high resolution, low cost, fast response, andability to operate reliably when the finger carries electrical noise, orwhen the touch surface is contaminated with dirt or moisture.

Because of the drawbacks of resistive devices, many attempts have beenmade to provide pointing capability based on capacitively sensing theposition of the finger. U.S. Pat. No. 3,921,166 to Volpe teaches acapacitive matrix in which the finger changes the transcapacitancebetween row and column electrodes. U.S. Pat. No. 4,103,252 to Bobickemploys four oscillating signals to interpolate x and y positionsbetween four capacitive electrodes. U. S. Pat. No. 4,455,452 to Schuylerteaches a capacitive tablet wherein the finger attenuates the capacitivecoupling between electrodes.

U.S. Pat. No. 4,550,221 to Mabusth teaches a capacitive tablet whereinthe effective capacitance to "virtual ground" is measured by anoscillating signal. Each row or column is polled sequentially, and arudimentary form of interpolation is applied to resolve the positionbetween two rows or columns. An attempt is made to address the problemof electrical interference by averaging over many cycles of theoscillating waveform. The problem of contamination is addressed bysensing when no finger was present, and applying a periodic calibrationduring such no-finger-present periods. U.S. Pat. No. 4,639,720 toRympalski teaches a tablet for sensing the position of a stylus. Thestylus alters the transcapacitance coupling between row and columnelectrodes, which are scanned sequentially. U.S. Pat. No. 4,736,191 toMatzke teaches a radial electrode arrangement under the space bar of akeyboard, to be activated by touching with a thumb. This patent teachesthe use of total touch capacitance, as an indication of the touchpressure, to control the velocity of cursor motion. Pulsed sequentialpolling is employed to address the effects of electrical interference.

U.S. Pat. Nos. 4,686,332 and 5,149,919, to Greanias, teaches a stylusand finger detection system meant to be mounted on a CRT. As a fingerdetection system, it's X/Y sensor matrix is used to locate the twomatrix wires carrying the maximum signal. With a coding scheme these twowires uniquely determine the location of the finger position to theresolution of the wire stepping. For stylus detection, Greanias firstcoarsely locates it, then develops a virtual dipole by driving all lineson one side of the object in one direction and all lines on the oppositeside in the opposite direction. This is done three times with differentdipole phases and signal polarities. Assuming a predetermined matrixresponse to the object, the three measurements present a set ofsimultaneous equations that can be solved for position.

U.S. Pat. No. 4,733,222 to Evans is the first to teach a capacitancetouch measurement system that interpolates to a high degree. Evansteaches a three terminal measurement system that uses a drive, sense andelectrode signal set (3 signals) in its matrix, and bases themeasurement on the attenuation effect of a finger on the electrode nodesignal (uses a capacitive divider phenomena). Evans sequentially scansthru each drive set to measure the capacitance. From the three largestresponses an interpolation routine is applied to determine fingerposition. Evans also teaches a zeroing technique that allows "no-finger"levels to be cancelled out as part of the measurement.

U.S. Pat. No. 5,016,008 to Gruaz describes a touch sensitive pad thatalso uses interpolation. Gruaz uses a drive and sense signal set (2signals) in the touch matrix and like Evans relies on the attenuationeffect of a finger to modulate the drive signal. The touch matrix issequentially scanned to read each matrix lines response. Aninterpolation program then selects the two largest adjacent signals inboth dimensions to determine the finger location, and ratiometricallydetermines the effective position from those 4 numbers.

Gerpheide, PCT application US90/04584, publication No. W091/03039,applies to a touch pad system a variation of the virtual dipole approachof Greanias. Gerpheide teaches the application of an oscillatingpotential of a given frequency and phase to all electrodes on one sideof the virtual dipole,and an oscillating potential of the same frequencyand opposite phase to those on the other side. Electronic circuitsdevelop a "balance signal" which is zero when no finger is present, andwhich has one polarity if a finger is on one side of the center of thevirtual dipole, and the opposite polarity if the finger is on theopposite side. To acquire the position of the finger initially, thevirtual dipole is scanned sequentially across the tablet. Once thefinger is located, it is "tracked" by moving the virtual dipole towardthe finger once the finger has moved more than one row or column.

Because the virtual dipole method operates by generating a balancesignal that is zero when the capacitance does not vary with distance, itonly senses the perimeter of the finger contact area, rather than theentire contact area. Because the method relies on synchronous detectionof the exciting signal, it must average for long periods to rejectelectrical interference, and hence it is slow. The averaging timerequired by this method, together with the necessity to searchsequentially for a new finger contact once a previous contact is lost,makes this method, like those before it, fall short of the requirementsfor a fast pointing device that is not affected by electricalinterference.

It should also be noted that all previous touch pad inventions that usedinterpolation placed rigorous design requirements on their sensing pad.Greanias and Evans use a complicated and expensive drive, sense andelectrode line scheme to develop their signal. Gruaz and Gerpheide use atwo signal drive and sense set. In the present invention the driving andsensing is done on the same line. This allows the row and columnsections to be symmetric and equivalent. This in turn allows independentcalibration of all signal paths, which makes board layout simpler andless constraining, and allows for more unique sensor topologies.

The shortcomings of the inventions and techniques described in the priorart can also be traced to the use of only one set of driving and sensingelectronics, which was multiplexed sequentially over the electrodes inthe tablet. This arrangement was cost effective in the days of discretecomponents, and avoided offset and scale differences among circuits.

The sequential scanning approach of previous systems also made them moresusceptible to noise. Noise levels could change between successivemeasurements, thus changing the measured signal and the assumptions usedin interpolation routines.

Finally, all previous approaches assumed a particular signal responsefor finger position versus matrix position. Because the transfer curveis very sensitive to many parameters and is not a smooth linear curve asGreanias and Gerpheide assume, such approaches are limited in the amountof interpolation they can perform.

It is thus an object of the present invention to provide atwo-dimensional capacitive sensing system equipped with a separate setof drive/sense electronics for each row and for each column of acapacitive tablet, wherein all row electrodes are sensed simultaneously,and all column electrodes are sensed simultaneously.

It is a further object of the present invention to provide an electronicsystem that is sensitive to the entire area of contact of a finger witha capacitive tablet, and to provide as output the coordinates of somemeasure of the center of this contact area while remaining insensitiveto the characteristic profile of the object being detected.

It is a further object of the present invention to provide an electronicsystem that provides as output some measure of area of contact of afinger with a capacitive tablet.

BRIEF DESCRIPTION OF THE INVENTION

With the advent of very high levels of integration, it has becomepossible to integrate many channels of driving/sensing electronics intoone integrated circuit, along with the control logic for operating them,and the interface electronics to allow the pointing device tocommunicate directly with a host microprocessor. The present inventionuses adaptive analog techniques to overcome offset and scale differencesbetween channels, and can thus sense either transcapacitance orself-capacitance of all tablet rows or columns in parallel. Thisparallel-sensing capability, made possible by providing one set ofelectronics per row or column, allows the sensing cycle to be extremelyshort, thus allowing fast response while still maintaining immunity tovery high levels of electrical interference.

The present invention comprises a position-sensing technologyparticularly useful for applications where finger position informationis needed, such as in computer "mouse" or trackball environments.However the position-sensing technology of the present invention hasmuch more general application than a computer mouse, because its sensorcan detect and report if one or more points are being touched. Inaddition, the detector can sense the pressure of the touch.

According to a preferred embodiment of the present invention, referredto herein as a "finger pointer" embodiment, a position sensing systemincludes a position sensing transducer comprising a touch-sensitivesurface disposed on a substrate, such as a printed circuit board,including a matrix of conductive lines. A first set of conductive linesruns in a first direction and is insulated from a second set ofconductive lines running in a second direction generally perpendicularto the first direction. An insulating layer is disposed over the firstand second sets of conductive lines. The insulating layer is thin enoughto promote significant capacitive coupling between a finger placed onits surface and the first and second sets of conductive lines.

Sensing electronics respond to the proximity of a finger to translatethe capacitance changes of the conductors caused by finger proximityinto position and touch pressure information. Its output is a simple X,Y and pressure value of the one object on its surface.

Different prior art pad scan techniques have different advantages indifferent environments. Parallel drive/sense techniques according to thepresent invention allow input samples to be taken simultaneously, thusall channels are affected by the same phase of an interfering electricalsignal, greatly simplifying the signal processing and noise filtering.

There are two drive/sense methods employed in the touch sensingtechnology of the present invention. According to a first and presentlypreferred embodiment of the invention, the voltages on all of the Xlines of the sensor matrix are simultaneously moved, while the voltagesof the Y lines are held at a constant voltage, with the complete set ofsampled points simultaneously giving a profile of the finger in the Xdimension. Next, the voltages on all of the Y lines of the sensor matrixare simultaneously moved, while the voltages of the X lines are held ata constant voltage to obtain complete set of sampled pointssimultaneously giving a profile of the finger in the other dimension.

According to a second drive/sense method, the voltages on all of the Xlines of the sensor matrix are simultaneously moved in a positivedirection, while the voltages of the Y lines are moved in a negativedirection. Next, the voltages on all of the X lines of the sensor matrixare simultaneously moved in a negative direction, while the voltages ofthe Y lines are moved in a positive direction. This technique doublesthe effect of any transcapacitance between the two dimensions, orconversely, halves the effect of any parasitic capacitance to ground. Inboth methods, the capacitive information from the sensing processprovides a profile of the proximity of the finger to the sensor in eachdimension.

Both embodiments then take these profiles and calculate the centroid forX and Y position and integrate under the curve for the Z pressureinformation. The position sensor of these embodiments can only reportthe position of one object on its sensor surface. If more than oneobject is present, the position sensor of this embodiment computes thecentroid position of the combined set of objects. However, unlike priorart, because the entire pad is being profiled, enough information isavailable to discern simple multi-finger gestures to allow for a morepowerful user interface.

According to another aspect of the present invention, several powerreduction techniques which can shut down the circuit betweenmeasurements have been integrated into the system. This is possiblebecause the parallel measurement technique according to the presentinvention is so much faster than prior art techniques.

According to a further aspect of the invention, noise reductiontechniques that are focused on reducing noise produced in typicalcomputer environments are integrated into the system.

According to yet another aspect of the present invention, a capacitancemeasurement technique which is easier to calibrate and implement isemployed.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1a is a top view of an object position sensor transducer accordingto a presently preferred embodiment of the invention showing the objectposition sensor surface layer including a top conductive trace layer andconductive pads connected to a bottom trace layer.

FIG. 1b is a bottom view of the object position sensor transducer ofFIG. 1 a showing the bottom conductive trace layer.

FIG. 1c is a composite view of the object position sensor transducer ofFIGS. 1a and 1b showing both the top and bottom conductive trace layers.

FIG. 1d is a cross-sectional view of the object position sensortransducer of FIGS. 1a-1c.

FIG. 2 is a block diagram of sensor decoding electronics which may beused with the sensor transducer in accordance with a preferredembodiment of the present invention.

FIG. 3a is a simplified schematic diagram of a charge integrator circuitwhich may be used in the present invention.

FIG. 3b is a schematic diagram of an illustrative schematic diagram ofthe charge integrator circuit of FIG. 3a.

FIG. 4 is a timing of the operation of charge integrator circuit ofFIGS. 3a and 3b.

FIG. 5 is a schematic diagram of an illustrative filter and sample/holdcircuit for use in the present invention.

FIG. 6a is a schematic diagram of an illustrative minimum selector andsubtractor circuit including peak rejection which may be employed in thepresent invention, showing circuit details of four individual channelsand their interconnection.

FIG. 6b is a representation of what the output of the minimum selectorand subtractor circuit of FIG. 6a would be like without the backgroundlevel removed.

FIG. 6c is a representation of the output of the minimum selector andsubtractor circuit of FIG. 6a with the background level removed.

FIG. 7 is a schematic diagram of an illustrative OTA circuit used in theminimum selector and subtractor circuit, showing how the outputs Poutand Zout are derived, and further showing a current sink and sourceoptions, Poutn and Poutp, respectively, for the Pout output.

FIG. 8 is a schematic diagram of an illustrative maximum detectorcircuit which may be used in the present invention.

FIG. 9a is a schematic diagram of an illustrative position encodercircuit which may be used in the present invention.

FIG. 9b is a schematic diagram of an P-type OTA circuit which may beused in the position encoder circuit of the present invention.

FIG. 9c is a schematic diagram of an N-type OTA circuit which may beused in the position encoder circuit of the present invention.

FIG. 10 is a schematic diagram of an illustrative ZSum circuit which maybe used in the present invention.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT

This application is a continuation-in-part of co-pending applicationSer. No. 07/895,934, filed Jun. 8, 1992. The present invention continuesthe approach disclosed in the parent application and provides moreunique features not previously available. These improvements provideincreased sensitivity, and greater noise rejection, increased dataacquisition rate and decreased power consumption.

Those of ordinary skill in the art will realize that the followingdescription of the present invention is illustrative only and not in anyway limiting. Other embodiments of the invention will readily suggestthemselves to such skilled persons.

The present invention brings together in combination a number of uniquefeatures which allow for new applications not before possible. Becausethe object position sensor of the present invention has very low powerrequirements, it is beneficial for use in battery operated or low powerapplications such as lap top or portable computers. It is also a verylow cost solution, has no moving parts (and is therefore virtuallymaintenance free), and uses the existing printed circuit board tracesfor sensors. The sensing technology of the present invention can beintegrated into a computer motherboard to even further lower its cost incomputer applications. Similarly, in other applications the sensor canbe part of an already existent circuit board.

Because of its small size and low profile, the sensor technology of thepresent invention is useful in lap top or portable applications wherevolume is important consideration. The sensor technology of the presentinvention requires circuit board space for only a single sensorinterface chip that can interface directly to a microprocessor, plus thearea needed on the printed circuit board for sensing.

The sensor material can be anything that allows creation of a conductiveX/Y matrix of pads. This includes not only standard PC board, but alsoflexible PC board, conductive elastomer materials, silk-screenedconductive lines, and piez-oelectric Kynar plastic materials. Thisrenders it useful as well in any portable equipment application or inhuman interface where the sensor needs to be molded to fit within thehand.

The sensor can be conformed to any three dimensional surface. Copper canbe plated in two layers on most any surface contour producing thesensor. This will allow the sensor to be adapted to the best ergonomicform needed for a application. This coupled with the "light-touch"feature will make it effortless to use in many applications. The sensorcan also be used in an indirect manner, i.e it can have a conductivefoam over the surface and be used to detect any object (not justconductive) that presses against it's surface.

Small sensor areas are practical, i.e., a presently conceived embodimenttakes about 1.5"×1.5" of area, however those of ordinary skill in theart will recognize that the area is scalable for different applications.The matrix area is scaleable by either varying the matrix trace spacingor by varying the number of traces. Large sensor areas are practicalwhere more information is needed.

Besides simple X and Y position information, the sensor technology ofthe present invention also provides finger pressure information. Thisadditional dimension of information may be used by programs to controlspecial features such as "brush-width" modes in Paint programs, specialmenu accesses, etc., allowing provision of a more natural sensory inputto computers. It has also been found useful for implementing "mouseclick and drag" modes and for simple input gestures.

The user will not even have to touch the surface to generate the minimumreaction. This feature can greatly minimize user strain and allow formore flexible use.

The sense system of the present invention depends on a transducer devicecapable of providing position and pressure information regarding theobject contacting the transducer. Referring first to FIGS. 1a-1d, top,bottom, composite, and cross-sectional views, respectively, are shown ofa presently-preferred touch sensor array for use in the presentinvention. Since capacitance is exploited by this embodiment of thepresent invention, the sensor surface is designed to maximize thecapacitive coupling.

A presently preferred sensor array 10 according to the present inventioncomprises a substrate 12 including a set of first conductive traces 14disposed on a top surface 16 thereof and run in a first direction tocomprise row positions of the array. A set of second conductive traces18 are disposed on a bottom surface 20 thereof and run in a seconddirection preferably orthogonal to the first direction to form thecolumn positions of the array. The sets of first and second conductivetraces 14 and 18 are alternately in contact with periodic sense pads 22comprising enlarged areas, shown as diamonds in FIGS. 1a-1c. While sensepads 22 are shown as diamonds in FIGS. 1a-1c, any shape, such ascircles, which allows close packing of the sense pads, is equivalent forpurposes of this invention. As an arbitrary convention herein, the setof first conductive traces 14 will be referred to as being oriented inthe "X" or "row" direction and may be referred to herein sometimes as "Xlines" and the set of second conductive traces 18 will be referred to asbeing oriented in the "Y" or "column" direction and may be referred toherein sometimes as "Y lines".

The number and spacing of these sense pads 22 depends upon theresolution desired. For example, in an actual embodiment constructedaccording to the principles of the present invention, a 0.10 inchcenter-to-center diamond-shaped pattern of sense pads disposed along amatrix of 15 rows and 15 columns of conductors is employed. Every othersense pad 22 in each direction in the pad pattern is connected sets offirst and second conductive traces 14 and 18 on the top and bottomsurfaces 16 and 20, respectively of substrate 12.

Substrate 12 may be a printed circuit board, a flexible circuit board orany of a number of available circuit interconnect technology structures.Its thickness is unimportant as long as contact may be made therethroughfrom the bottom conductive traces 18 to their sense pads 22 on the topsurface 16. The printed circuit board comprising substrate 12 can beconstructed using standard industry techniques. Board thickness is notimportant. Connections from the sense pads 22 to the bottom traces 18may be made employing standard plated-through hole techniques well knownin the printed circuit board art.

In an alternate embodiment of the present invention, the substratematerial 12 may have a thickness on the order of 0.005 to 0.010 inches.Then the diamonds on the top surface 16 and the plated thru holes thatconnect to the bottom surface traces 18, can be omitted, furtherreducing the cost of the system.

An insulating layer 24 is disposed over the sense pads 22 on top surface16 to insulate a human finger or other object therefrom. Insulatinglayer 24 is preferably a thin layer (i.e., approximately 5 mils) to keepcapacitive coupling large and may comprise a material, such as mylar,chosen for its protective and ergonomic characteristics. The term"significant capacitive coupling" as used herein shall mean capacitivecoupling having a magnitude greater than about 0.5 pF.

There are two different capacitive effects taking place when a fingerapproaches the sensor array 10. The first capacitive effect istrans-capacitance, or coupling between sense pads 22, and the secondcapacitive effect is self-capacitance, or coupling to virtual ground.Sensing circuitry is coupled to the sensor array 10 of the presentinvention and responds to changes in either or both of thesecapacitances. This is important because the relative sizes of the twocapacitances change greatly depending on the user environment. Theability of the present invention to detect changes in both selfcapacitance and trans-capacitance results in a very versatile systemhaving a wide range of applications.

According to the preferred embodiment of the invention, a positionsensor system including sensor array 10 and associated touch detectorcircuitry will detect a finger position on a matrix of printed circuitboard traces via the capacitive effect of finger proximity to the sensorarray 10. The position sensor system will report the X, Y position of afinger placed near the sensor array 10 to much finer resolution than thespacing between the row and column traces 14 and 18. The position sensoraccording to this embodiment of the invention will also report a Z valueproportional to the outline of that finger and hence indicative of thepressure with which the finger contacts the surface of insulating layer24 over the sensor array

According to the presently preferred embodiment of the invention, a verysensitive, light-touch detector circuit may be provided using adaptiveanalog VLSI techniques. The circuit of the present invention is veryrobust and calibrates out process and systematic errors. The detectorcircuit of the present invention will process the capacitive inputinformation and provide digital information to a microprocessor.

According to this embodiment of the invention, sensing circuitry iscontained on a single sensor processor integrated circuit chip. Thesensor processor chip can have any number of X and Y "matrix" inputs.The number of X and Y inputs does not have to be equal. The Integratedcircuit has a digital bus as output. In the illustrative exampledisclosed in FIGS. 1a-1d herein, the sensor array 10 has 15 traces inboth the Y and Y directions. The sensor processor chip thus has 15 Xinputs and 15 Y inputs.

The X and Y matrix nodes are driven and sensed in parallel, with thecapacitive information from each line indicating how close a finger isto that node. The scanned information provides a profile of the fingerproximity in each dimension. According to this aspect of the presentinvention, the profile centroid is derived in both the X and Ydirections and is the position in that dimension. The profile curve ofproximity is also integrated to provide the Z information.

There are two drive and sense methods employed in the touch sensingtechnology of the present invention. According to a first and presentlypreferred embodiment of the invention, the voltages on all of the Xlines of the sensor array 10 are simultaneously moved, while thevoltages of the Y lines are held at a constant voltage. Next, thevoltages on all of the Y lines of the sensor array 10 are simultaneouslymoved, while the voltages of the X lines are held at a constant voltage.This scanning method accentuates the measurement of capacitance tovirtual ground provided by the finger. Those of ordinary skill in theart will recognize that order of these two steps is somewhat arbitraryand may be reversed.

According to a second drive/sense method, the voltages on all of the Xlines of the sensor array 10 are simultaneously moved in a positivedirection, while the voltages of the Y lines are moved in a negativedirection. Next, the voltages on all of the X lines of the sensor array10 are simultaneously moved in a negative direction, while the voltagesof the Y lines are moved in a positive direction. This seconddrive/sense method accentuates transcapacitance and de-emphasizesvirtual ground capacitance. As with the first drive/sense method, thoseof ordinary skill in the art will recognize that order of these twosteps is somewhat arbitrary and may be reversed.

Referring now to FIG. 2, a block diagram of the presently preferredsensing circuitry 30 for use according to the present invention ispresented. This block diagram shows the sensing circuitry 30 in onedimension (X) only. Those of ordinary skill in the art will appreciatethat an identical circuit would be used for sensing the opposite (Y)dimension. Such skilled persons will further note that these dimensionsdo not need to be orthogonal. For example, they can be radial or of anyother nature to match the contour of the sensing pad and the needs ofthe system.

The capacitance at each sensor matrix node is measured simultaneouslyusing charge integrator circuits 32-1 through 32-n. The function of eachcharge integrator is to develop an output voltage proportional to thecapacitance sensed on the corresponding X matrix line.

According to the presently preferred drive/sense method, the capacitancemeasurements are performed simultaneously across all inputs in onedimension to overcome a problem which is inherent in all prior artapproaches that scan individual inputs. The problem with the prior-artapproach is that it is sensitive to high frequency and large amplitudenoise (large dv/dt noise) that is coupled to the circuit via thetouching object. Such noise may distort the finger profile because ofnoise appearing in a later scan cycle but not an earlier one, due to achange in the noise level. The present invention overcomes this problemby taking a snapshot of all inputs simultaneously. The injected noise isproportional to the finger signal strength across all inputs andtherefore symmetric around the finger centroid. Because it is symmetricaround the finger centroid it does not affect the finger position.

Because of the nature of the charge integrator circuits 32-1 through32-n, their outputs will be changing over time and will have the desiredvoltage output for only a short time. This desired voltage is capturedby the filter and sample/hold circuits 34-1 through 34-n. As controlledby the control circuitry, 36, the filter and sample/hold circuits 34-1through 34-n will capture the desired voltage and store it.Additionally, the result may also be filtered depending on the size ofthe sample and hold capacitor in the cell.

The filter and sample/hold circuits 34-1 through 34-n then provides aninput for the Minimum Selector and Subtractor circuit 38, which computesan average of its n smallest input values (n=3 is presently preferred)and subtracts that value from each input. Minimum Selector andSubtractor circuit 38 then generates a current output for every inputwhich is proportional to the difference at that input between the actualvalue at the input and the computed average minimum value. This circuitperforms the task of subtracting out the background capacitance seen bythe sensing circuitry and then providing a current proportional to theadditional capacitance seen above the background level. If n=1, then theminimum value is selected. Any n>1 will select an average of the nvalues.

This current is then replicated and sent to two destinations. One copyis sent to the position encoder circuit 40, and the second is sent tothe Zsum circuit 42.

The position encoder circuit 40 uses the current inputs as weights, andprovides a scaled weighted mean (centroid) of the set of input currentsand their relation to their position in the sensor. Position encodercircuit 40 is a linear position encoder having a voltage output whichvaries between the power supply rails. Because the circuit produces acontinuous weighted mean over all input values, it is capable ofinterpolation to a much finer resolution than the spacing of the matrixgrid spacing.

The output of the position encoder circuit 40 is then presented tosample/hold circuit 44-1 and analog to digital (ND) converter 46-1. Theoperation of this portion of the circuit uses devices that are wellknown to those knowledgeable in the art.

The minimum selector and subtractor circuit 38 also generates a secondset of outputs which are all tied together and sent to the Zsum circuit42. Since these lines are shorted together, the individual outputcurrents are all summed together. The total effect of the finger on thesensor in one dimension is thus integrated, producing a current sumresult proportional to the pressure or proximity of the input object.The Zsum circuit 42 takes this current sum and then converts it back toa voltage which is proportional to the current sum. Those of ordinaryskill in the art will appreciate that there are many conversion choicesdepending on the particular use to which the invention is put, it can bea linear conversion, square root (compressive), or squared (expansive).In presently preferred embodiment a compressive conversion is chosen tocompress large changes and emphasize small changes, since detecting verylight touches is of particular interest.

The output of the ZSum circuit 42 is presented to a sample/hold circuit44-2 which stores the results. The output of sample hold circuit 44-2drives A/D converter circuit 46-2 which converts the analog informationto a digital form useable by microcomputers.

Control circuitry 36 of FIG. 2 orchestrates the operation of theremainder of the circuitry. Because the system is discretely sampled andpipelined in its operation, control circuitry 36 is present to managethe signal flow. The functions performed by control circuitry 36 may beconventionally developed via what is commonly known in the art as astate machine or by a microcontroller.

The output of the filter and sample/hold circuits, 34-1 thru 34-n, isalso monitored by a maximum detector circuit 47. The purpose of thissection of circuitry is to generate an interrupt signal to amicroprocessor if there is a finger signal greater than a presetthreshold. The maximum detector circuit 47 outputs a signal which isrelated to the largest input voltage. This signal is then comparedagainst a predetermined threshold, noted as VTHMAX with the comparator48. If the signal is greater than the preset threshold the comparatorwill output a logic 1 level which, after conditioned with proper timingthru AND gate 49, provides an interrupt signal to a microprocessor.Those skilled in the art will recognize that this signal is not limitedto being an interrupt and could be used for polling, for example, or inother ways that better fit the needs of the entire system.

The structure and operation of the individual blocks of FIG. 2 will nowbe disclosed. Referring now to FIGS. 3a, 3b, and 4, a typical chargeintegrator circuit will be described. Charge integrator circuit 32 isshown as a simplified schematic diagram in FIG. 3a and as anillustrative schematic diagram in FIG. 3b. The timing of the operationof charge integrator circuit 32 is shown in FIG. 4.

Charge integrator circuit 32 is based on the fundamental physicalphenomena of using a current to charge a capacitor. If the capacitor ischarged for a constant time by a constant current then a voltage will beproduced on the capacitor which is inversely proportional to thecapacitance. The capacitance to be charged is the sensor array linecapacitance in parallel with an internal capacitor. This internalcapacitor will contain the voltage of interest.

Referring now to FIG. 3a, a simplified schematic diagram of anillustrative charge integrator circuit 32 is shown. A charge integratorcircuit input node 50 is connected to one of the X (or Y) lines of thesensor array. A first shorting switch 52 is connected between the chargeintegrator circuit input node 50 and V_(DD), the positive supply rail. Asecond shorting switch 54 is connected between the charge integratorcircuit input node 50 and ground, the negative supply rail. A positiveconstant current source 56 is connected to V_(DD), the positive supplyrail and to the charge integrator circuit input node 50 and through afirst current source switch 58. A negative constant current source 60 isconnected to ground and to the charge integrator circuit input node 50and through a second current source switch 62.

A first internal capacitor 64 is connected between V_(DD) and outputnode 66 of charge integrator circuit 32. A positive voltage storageswitch 68 is connected between output node 66 and input node 50. Asecond internal capacitor 70 has one of its plates connected to groundthrough a switch 72 and to output node 66 of charge integrator circuit32 through a switch 74, and the other one of its plates connected toinput node 50 through a negative voltage storage switch 76 and to V_(DD)through a switch 78. The capacitance of first and second internalcapacitors 64 and 70 should be a small fraction (i.e., about 10%) of thecapacitance of the individual sensor array lines. In a typicalembodiment, the sensor array line capacitance will be about 10 pF andthe capacitance of capacitors 64 and 70 should be about 1 pF.

According to the presently preferred embodiment of the invention, theapproach used is a differential measurement for added noise immunity,the benefit of which is that any low frequency common mode noise getssubtracted out. For the following discussion, it is to be assumed thatall switches are open unless they are noted as closed. First, the sensorarray line is momentarily shoaled to V_(DD) through switch 52, switch 68is closed connecting capacitor 64 in parallel with the capacitance ofthe sensor line. Then the parallel capacitor combination is dischargedwith a constant current from current source 60 through switch 62 for afixed time period. At the end of the fixed time period, switch 68 isopened, thus storing the voltage on the sensor array line on capacitor64.

The sensor line is then momentarily shorted to ground through switch 54,and switches 72 and 76 are closed to place capacitor 70 in parallel withthe capacitance of the sensor line. Switch 58 is closed and the parallelcapacitor combination is charged with a constant current from currentsource 56 for a fixed time period equal to the fixed time period of thefirst cycle. At the end of the fixed time period, switch 76 is opened,thus storing the voltage on the sensor array line on capacitor 70.

The first and second measured voltages are then averaged. This isaccomplished by opening switch 72 and closing switches 78 and 74, whichplaces capacitor 70 in parallel with capacitor 64. Because capacitors 64and 70 have the same capacitance, the resulting voltage across them isequal to the average of the voltages across each individually. Thisfinal result is the value that is then passed onto the appropriate oneof filter and sample/hold circuits 34-1 through 34-n.

The low frequency noise, notably 50/60 Hz and their harmonics, behavesas a DC current component that adds in one measurement and subtracts inthe other. When the two results are added together that noise componentis canceled out. The amount of noise rejection is a function of howquickly in succession the two charge/discharge cycles are performed. Oneof the reasons for the choice of this charge integrator circuit is thatit allows measurements to be taken quickly.

Referring now to FIG. 3b, a more complete schematic diagram of anillustrative embodiment of charge integrator circuit 32 of thesimplified diagram of FIG. 3a is shown. Input node 50 is shown connectedto V_(DD) and ground through pass gates 80 and 82, which replaceswitches 52 and 54 of FIG. 3a. Pass gate 80 is controlled by a signalResetUp presented to its control input and pass gate 82 is controlled bya signal ResetDn presented to its control input. Those of ordinary skillin the art will recognize that pass gates 80 and 82, as well as all ofthe other pass gates which are represented by the same symbol in FIG. 3bmay be conventional CMOS pass gates as are known in the art. Theconvention used herein is that the pass gate will be off when itscontrol input is held low and will be on and present a low impedanceconnection when its control input is held high.

P-Channel MOS transistors 84 and 86 are configured as a current mirror.P-Channel MOS transistor 84 serves as the current source 56 and passgate 88 serves as switch 58 of FIG. 3a. The control input of pass gate88 is controlled by a signal StepUp,

N-Channel MOS transistors 92 and 94 are also configured as a currentmirror. N-Channel MOS transistor 92 serves as the current source 60 andpass gate 96 serves as switch 62 of FIG. 3a. The control input of passgate 96 is controlled by a signal StepDn. P-Channel MOS transistor 90and N-Channel MOS transistor 98 are placed in series with P-Channel MOScurrent mirror transistor 86 and N-Channel MOS current mirror transistor94. The control gate of P-Channel MOS transistor 90 is driven by anenable signal EN, which turns on P-Channel MOS transistor 90 to energizethe current mirrors. This device is used as a power conservation deviceso that the charge integrator circuit 32 may be turned off to conservepower when it is not in use.

N-Channel MOS transistor 98 has its gate driven by a reference voltageVref, which sets the current through current mirror transistors 86 and94. The voltage Vref may be individually adjusted for each chargeintegrator circuit 32 to compensate for manufacturing variations. EachVref may be developed from a analog programmable voltage source, such asdescribed in van Steenwijk, Hoen, and Wallinga "A Nonvolatile AnalogVoltage Programmable Voltage Source Using VIP MOS EEPROM Structure,"IEEE Journal of Solid State Circuits, Jul. 1993. Alternatively, awritable analog reference voltage storage device, such as disclosed inU.S. Pat. No. 5,166,562 may be employed. This allows the circuit to becalibrated in the factory to zero out process variations as well ascapacitance variations in the sensor. The calibration focus is togenerate a constant and equal output from all the charge integratorcircuits 32-1 through 32-n of FIG. 2 if no finger is present. Althoughthe present approach is very robust, those of ordinary skill in the artwill also appreciate an embodiment in which calibration will be allowedto occur in real time (via long time constant feedback) thereby zeroingout any long term effects due to sensor environmental changes.

Note that proper sizing of MOS transistors 94 and 98 may providetemperature compensation. This is accomplished by taking advantage ofthe fact that the threshold of N-Channel MOS transistor 98 reduces withtemperature while the mobility of both N-Channel MOS transistors 94 and98 reduce with temperature. The threshold reduction has the effect ofincreasing the current while the mobility reduction has the effect ofdecreasing the current. By proper device sizing these effects can canceleach other out over a significant part of the operating range.

Capacitor 64 has one plate connected to V_(DD) and the other plateconnected to the output node 66 and to the input node 50 through passgate 100, shown as switch 68 in FIG. 3a. The control input of pass gate100 is driven by the control signal SUp. One plate of capacitor 70 isconnected to input node 50 through pass gate 102 (switch 76 in FIG. 3a)and to VDD through pass gate 104 (switch 72 in FIG. 3a). The controlinput of pass gate 102 is driven by the control signal SDn and thecontrol input of pass gate 104 is driven by the control signal ChUp. Theother plate of capacitor 70 is connected to ground through N-Channel MOStransistor 106 (switch 72 in FIG. 3a) and to output node 66 through passgate 108. The control input of pass gate 108 is driven by control signalShare.

Referring now to FIGS. 3a, 3b and the timing diagram of FIG. 4, theoperation of charge integrator circuit 32 during one scan cycle may beobserved. First the EN (enable) control signal goes active by going to 0v. This turns on the current mirrors and energizes the charge anddischarge current sources, MOS transistors 84 and 92. The ResetUpcontrol signal is active high at this time, which shorts the input node50 (and the sensor line to which it is connected) to V_(DD). The SUpcontrol signal is also active high at this time which connects capacitor64 and the output node 66 to input node 50. This arrangement guaranteesthat the following discharge portion of the operating cycle alwaysstarts from a known equilibrium state.

The discharge process starts after ResetUp control signal goes inactive.The StepDn control signal goes active, connecting MOS transistor 92, thedischarge current source, to the input node 50 and its associated sensorline. StepDn is active for a set amount of time, allowing the combinedcapacitance of the sensor line and capacitor 64 to charge down duringthat time. StepDn is then turned off. A short time later the SUp controlsignal goes inactive, storing the measured voltage on capacitor 64 toend the discharge cycle.

Next, ResetDn control signal becomes active and shorts the sensor lineto ground. Simultaneously the SDn and ChDn control signals become activeand connect capacitor 70 between ground and the sensor line. Capacitor70 is discharged to ground, guaranteeing that the following charge upcycle always starts from a known state.

The chargeup cycle starts after ResetDn control signal becomes inactiveand the StepUp control signal becomes active. At this point the currentcharging source, MOS transistor 84, is connected to the sensor line andthe sensor line charges up. The StepUp control signal is active for aset amount of time (preferably equal to the time for the previouslymentioned cycle) allowing the capacitance to charge, and then it isturned off. The SDn control signal then goes inactive, leaving themeasured voltage across capacitor 70.

The averaging cycle now starts. First the voltage on capacitor 70 islevel shifted. This is done by the ChDn control signal going inactive,letting one plate of the capacitor 70 float. Then the ChUp controlsignal goes active, connecting the second plate of the capacitor toV_(DD). Then the Share control signal becomes active which connects thefirst plate of capacitor 70 to output node 66, thus placing capacitors64 and 70 in parallel. This has the effect of averaging the voltagesacross the two capacitors, thus subtracting out common mode noise aspreviously described. This average voltage is also then available onoutput node 66.

According to the present invention, two different drive/sense methodshave been disclosed. Those of ordinary skill in the art will readilyobserve that the charge integrator circuit 32 disclosed with referenceto FIGS. 3a, 3b, and 4 is adaptable to operate according to eitherscanning method disclosed herein.

As is clear from an understanding of the operation of charge integratorcircuit 32, its output voltage is only available for a short period oftime. In order to capture this voltage a sample/hold circuit is used.Referring now to FIG. 5, a schematic diagram of an illustrative filterand sample/hold circuit is presented. Those of ordinary skill in the artwill recognize this circuit, which comprises an input node 112, a passgate 114 having a control input driven by a Sample control signal, acapacitor 116 connected between the output of the pass gate 114 and afixed voltage such as ground, and an output node comprising the commonconnection between the capacitor 116 and the output of the pass gate114. In a typical embodiment, capacitor 116 will have a capacitance ofabout 10 pF.

The sample/hold circuit of FIG. 5 is well known in the art, but isapplied in a way so that it acts as a filter as well. The filter timeconstant is K times the sample signal period, where K is the ratio ofcapacitor 116 to the sum of capacitors 64 and 70 of the chargeintegrator circuit 32 of FIGS. 3a and 3b. This filter further reducesnoise injection. In the preferred embodiment, K=10/2=5.

As shown in FIG. 2, the output of all of the filter and sample/holdcircuits 34-1 thru 34-n drive minimum selector and subtractor circuit38. Referring now to FIG. 6a, a schematic diagram of an illustrativeminimum selector and subtractor circuit 38 useful for employment in thepresent invention is shown. The illustrative circuit of FIG. 6a is shownhaving four channels, although those of ordinary skill in the art willreadily recognize that the circuit could be arbitrarily extended to alarger number of channels.

The minimum selector and subtractor circuit 38 is designed to take a setof inputs, detect the average of the three smallest input values andsubtract that average value from each individual value in the entireinput set. The circuit then produces a current which is proportional tothis subtracted value, which for most background inputs will be zero.This sequence of steps is illustrated in FIGS. 6b and 6c for an examplewhere there are 15 inputs (X1 to X15) in the set. FIG. 6b shows theinput to the minimum selector/subtractor circuit as generated by theFilter circuits, 34-1 thru 34-n. The drawing shows a typical fingerprofile with the background or minimum level noted. After the minimumselector and subtractor circuit 38 processes the input, it produces anoutput like that shown in FIG. 6c, which is the input set with thebackground value subtracted out.

Each individual channel, even though constituting a single functionalunit, can be thought of as consisting of a minimum selector circuit 120and a subtractor circuit 122. In the minimum selector circuit 120, theactive elements are P-channel MOS transistors 124a-124d, each having itssource electrode connected to an intermediate node 126a-126d, its gateelectrode connected to the channel input nodes 128a-128d, its drainelectrode connected to the drain electrodes of N-channel MOScurrent-limiting transistors 130a-130d, respectively. N-channel MOStransistors 130a-130d have their source electrodes connected to a fixedvoltage, such as ground, and their gate electrodes held at a potentialV_(sink) above (more positive than) their source electrodes such thatthey function as current sinks with limited voltage compliance.

Each intermediate node 126a-126d is also connected to a current source132a-132d that supplies the operating current of the P-channel MOStransistors 124a-124d from a fixed voltage source V_(DD). Intermediatenodes 126a-126d are also connected to the non-inverting input of anoperational transconductance amplifier 134a-134d (OTA), which comprisesthe heart of the subtractor circuit 122 of each channel. A pass gate136a-136d allows connecting the intermediate node 126a-126d to a minimumrail 138 which is common to all signal channels in the system. Theinverting input of each OTA 134a-134d is connected to a storagecapacitor 140a-140d as well as to one end of a pass gate 142a-142d whichpermits selectively connecting the inverting input of the OTA 134a-134dto its Pout output 144a-144d. Each OTA also has a Zout output 146a-146d.

The N-Channel MOS current sink transistors 130a-130d have the effect oflimiting the current that any one of the P-channel MOS transistors124a-124d can draw. As is well known to those of ordinary skill in theart, this is because the common drain connections of the N-channel andP-channel MOS transistors will assume such a potential as to reduce thedrain-to-source voltage difference of one or the other of the twotransistors far enough to prevent it from drawing more current than theother transistor allows to flow.

If the sink current of N-Channel MOS current sink transistors 130a-130dis chosen to be larger than the current of sources 132a-132d but smallerthan the sum of all source currents, then no single P-Channel MOStransistor 124a-124d can conduct all the current in the minimumselection phase of operation. Instead, several transistors have to shareit. Thus it may be seen that the minimum select and subtractor circuitof FIG. 6a selects not the absolute minimum of the input voltages of allchannels, but rather an average of the several lowest input voltagesfrom among all channels. The gate voltage of the N-Channel MOScurrent-limiting transistors 130a-130d is selected to set theirsaturation current to be W/n times the value of the current source 132,where W is the number of channels present in the system and n is thenumber of channels to be averaged over to obtain the minimum.

For example, assuming an embodiment including fifteen identical signalchannels, and the currents of the N-channel MOS transistors 130 in eachchannel are five times larger than the currents of current sources 132,then the three P-channel transistors 124 with the lowest gate potentialsmust share the total system current because the total is equal tofifteen times the source current. The effect is to reject or at leastattenuate negative peaks on the input voltages. This property is highlydesirable in applications where it is expected that several or mostchannel input voltages will always be equal to a common minimum orbaseline potential, but that considerable noise existing on the inputsmay create false negative peaks.

As may be seen from FIG. 6a, each OTA 134a-134d has two output types,designated as Pout 144a-144d and Zout 146a-146d. Referring now to FIG.7, the circuitry for generating these outputs may be seen in detail. Asshown in FIG. 7, each OTA comprises N-Channel MOS input transistors 148and 150, P-Channel MOS current mirror pairs configured from transistors152 and 154, 156 and 158, and N-Channel MOS bias transistor 160. AnN-Channel MOS current mirror comprising transistors 162 and 164 isconnected to P-Channel current mirror transistors 152 and 158 as shown.

As thus far described, the circuit is conventional and the common drainnode of transistors 158 and 164 would form an output node for thecircuit. An extra P-channel MOS transistor 166 and N-Channel MOStransistor 168 are added to the circuit to form a second output node byreplicating the output buffer of the typical wide range outputtransamplifier. The common drain node of transistors 166 and 168,through pass gate 170, forms the Pout output section of the circuit. Thediode-connected N-Channel MOS transistor 172 between the common drainnode of transistors 158 and 164 and the Zout node assures that the Zoutlines of minimum selector and subtractor circuit 38 will only sourcecurrent, thus guaranteeing that only object-created signals above thebaseline contribute to the Zout signal.

In the presently preferred embodiment of the invention, the Pout outputsof half of the minimum selector and subtractor circuit 38 are configuredas current source outputs and may be designated Poutp outputs. The otherhalf of the outputs are current sink outputs and may be designated Poutnoutputs. This feature is also shown in FIG. 7. Pass gate 170, controlledby control signal PosEn is present to disconnect the position encodeload during the sample phase of the minimum select and subtract circuit.The current source node for the Poutp outputs is at the output of passgate 170. In this configuration, transistors 174 and 176 are not presentand Poutp goes to output Pout. This output would be used to drive theBiasIn line of the position encoder OTA as shown in FIG. 9b. The currentsink path is developed from the transamplifier output current at theoutput of pass gate 170. That current is fed into an NMOS current mirrorcomprising N-Channel MOS transistors 174 and 176. The drain of N-Channeltransistor 176 is the current sink Poutn output node and is connected tooutput Pout. This output would be used to drive the BiasIn line of theposition encoder OTA as shown in FIG. 9c.

Referring now to FIG. 8, an illustrative maximum detector circuit 47 foruse in the present invention is shown. As previously mentioned, thefunction of maximum detector circuit 47 is to monitor the outputs offilter and sample/hold circuits, 34-1 thru 34-n and to generate aninterrupt signal to a microprocessor if there a finger signal greaterthan a preset threshold VTHMAX is present. Those skilled in the art willrecognize that the signal is not limited to being an interrupt and couldbe used for other purposes such as polling etc.

Maximum detector circuit 47 includes an N-channel MOS bias transistor182 having its source connected to ground and its gate connected to abias voltage V_(BIAS) at node 184. The inputs of the maximum detectorcircuit 47 are connected to the outputs of filter and sample/holdcircuits 34-1 to 34-n as shown in FIG. 2. In the maximum detectorcircuit 47 illustrated in FIG. 8, there are (n) inputs. Each inputsection comprises a series pair of MOS transistors connected between thedrain of N-channel MOS bias transistor 182 and a voltage source V_(DD).

Thus, the input section for In₁ comprises P-channel MOS current-limitingtransistor 186 having its source connected to V_(DD) and its drainconnected to the drain of N-channel MOS input transistor 188. The gateof N-channel MOS input transistor 188 is connected to In₁ input node 190and the gate of P-channel MOS current-limiting transistor 186 isconnected to a source of bias voltage VLBIAS at node 192.

Similarly, the input section for In₂ comprises P-channel MOScurrent-limiting transistor 194 having its source connected to V_(DD)and its drain connected to the drain of N-channel MOS input transistor196. The gate of N-channel MOS input transistor 196 is connected to In₂input node 198 and the gate of P-channel MOS current-limiting transistor194 is connected to node 192.

The input section for In₃ comprises P-channel MOS current-limitingtransistor 200 having its source connected to VDD and its drainconnected to the drain of N-channel MOS input transistor 202. The gateof N-channel MOS input transistor 202 is connected to In₃ input node 204and the gate of P-channel MOS current-limiting transistor 200 isconnected to node 192.

The input section for In.sub.(n) comprises P-channel MOScurrent-limiting transistor 206 having its source connected to VDD andits drain connected to the drain of N-channel MOS input transistor 208.The gate of N-channel MOS input transistor 208 is connected toIn.sub.(n) input node 210 and the gate of N-channel MOS current-limitingtransistor 206 is connected to node 192. The sources of N-channel MOSinput transistors 188, 196, 202, and 208 are connected together to thedrain of N-channel MOS bias transistor 182. The output of maximumdetector circuit 47 is node 212 at the common connection of the drain ofN-channel MOS bias transistor 182 and the sources of the N-channel MOSinput transistors 188, 196, 202 and 208.

The maximum detector circuit 47 acts analogously to the minimum detectorcircuit 46 of parent application Ser. No. 07/895,934, filed Jun. 8,1992. The difference is that an N-channel bias transistor is usedinstead of a P-channel bias transistor and an N-channel transconductanceamplifier is used in place of a P-channel transconductance amplifier.The result is the output will now track approximately an N-channel biasdrop below the largest input (in non-averaging mode), since that muchdifference is needed to guarantee at least one input pair is on(186/188, 194/196 , . . . 206/208).

However for this circuit the output is not used for feedback, but isinstead used to drive a comparator 48 (FIG. 2) which is set to trip ifthe input is greater than the voltage V_(Thmax). If tripped, a MAXINTERRUPT signal is generated. The MAX INTERRUPT is used to "wake-up" amicroprocessor and tell it that there is an object detected at thesensor. The signal is prevented from appearing on the MAX INTERRUPT lineby AND gate 49 and a control signal from control circuitry 36. Thecontrol signal only allows the interrupt signal to pass after thecircuit has settled completely. The control signal presented to AND gate49 may be a SAMPLE signal which may be generated, for example, by thetrailing edge of the SHARE signal shown in FIG. 4.

As may be seen from FIG. 2, the outputs of minimum selector andsubtractor circuit 38 are presented to position encoder circuit 40.There are two identical position encoder circuits, one each for the Xand Y directions. The function of position encode circuit 40 is toconvert the input information into a signal representing objectproximity in the X (or Y) dimension of the sensor array matrix.According to a presently preferred embodiment of the invention, thiscircuit will provide a scaled weighted mean (centroid) of the set ofinput currents. The result is a circuit which is a linear positionencoder, having an output voltage which varies between the power supplyrails. Because it is a weighted mean, it averages all current inputs andcan in turn generate an output voltage which represents an X (or Y)position with a finer resolution than the spacing of the matrix gridspacing.

Referring now to FIG. 9a, a presently preferred embodiment of a positionencoder circuit 40 of FIG. 2 is shown in schematic diagram form. Becausethe position encoder circuits in the X and Y dimensions are identical,only one will be shown. The position encoder circuit 40 of FIG. 9a isshown having six inputs, but those of ordinary skill in the art willrecognize that, due to its symmetry, it may be arbitrarily expanded forother numbers of inputs.

As presently preferred, position encoder circuit 38 includes a pluralityof transconductance amplifiers 220-1 through 220-6 connected asfollowers. The outputs of all amplifiers 220-1 through 220-6 areconnected together to a common node 222, which comprises the positionencoder output node of the circuit 38.

The non-inverting inputs of amplifiers 220-1 through 220-6 are connectedto a resistive voltage divider network comprising resistors 224, 226,228, 230, 232, 234, and 236, shown connected between V_(DD) and ground.

Amplifiers 220-1 through 220-3 have P-channel MOS bias transistors anddifferential pair inputs due to the input operating range between zerovolts and V_(DD) /2, and are shown in schematic diagram form in FIG. 9b.The P-Channel MOS devices 250 and 252 form the differential input pairwhile 254 and 256 form a current mirror load. This is the standardconfiguration for a typical transconductance amplifier. Normally thebias current is provided by a P-Channel MOS current source device atnode 258. However in this application the bias current is providedexternally by the Poutp output (output of pass gate 170 in FIG. 7.) ofthe minimum selector/subtractor circuit through nodes 238, 240 and 242(I_(IN1) thru I_(IN3), respectively, of FIG. 9a).

Amplifiers 220-4 through 220-6 have N-channel MOS bias transistors anddifferential pair inputs due to the input operating range between V_(DD)/2 and V_(DD), and are shown in schematic diagram form in FIG. 9c. TheN-Channel MOS transistors 260 and 262 form the differential input pairwhile P-Channel MOS transistors 264 and 266 form a current mirror load.This is the standard configuration for a typical transconductanceamplifier. Normally the bias current is provided by a N-Channel MOScurrent source at node 268. However in this application the bias currentis provided externally by the Poutn output (drain of transistor 174 inFIG. 7) of the minimum selector and subtractor circuit through nodes244, 246 and 248 (I_(IN4) thru I_(IN6), respectively, of FIG. 9a). Thoseof ordinary skill in the art will readily recognize that amplifiers220-4 through 220-6 will be configured exactly like amplifiers 220-1through 220-3, except that all transistor and supply voltage polaritiesare reversed.

The position encoder circuit of FIG. 9a will provide a weighted mean(centroid) of the input currents weighted by the voltages on theresistor divider circuit to which the inputs of the amplifiers 220-1through 200-6 are connected. If the resistors 224, 226, 228, 230, 232,234, and 236 are all equal then the result is a circuit which is alinear position encoder, with its output voltage varying between thepower supply rails. Because it is a weighted mean, it averages allcurrent inputs which in turn generates an interpolated output. Thisarrangement affords finer resolution than the voltage spacing of voltagenodes "n" at the input. This is key to making a dense circuit function.This circuit is an improvement of a circuit described in DeWeerth,Stephen P., Analog VLSI Circuits For Sensorimotor Feedback, Ph.D Thesis,California Institute of Technology, 1991.

The output voltage of X position encoder circuit 40 is presented tosample/hold circuit 44-1, the output of which, as is well known in theart, either follows the input or holds a value present at the inputdepending on the state of its control input. The structure and operationof sample/hold circuits are well known in the art.

The output of sample/hold circuit 44-1 drives the input ofanalog-to-digital (ND) converter 46-1. The output of A/D converter 46-1is a digital value proportional to the position of the object in the Xdimension of the sensor array matrix 10.

Referring now to FIG. 10, a schematic diagram of a presently preferredembodiment of a ZSum circuit 42 is shown. ZSum circuit 42 takes acurrent as an input on node 270. If N-Channel MOS transistor 272 was notpresent then the combination of N-Channel MOS transistors 274 and 276would be a current mirror and the current on input node 270 would appearon node 278. Transistor 272 is a source degradation resistor that,depending on the gain setting on node 280, reduces the current mirrortransfer factor from an ideal factor of 1 to something less than 1. Thesmaller the voltage present on node 280 the smaller the transfer factor.

P-Channel MOS transistors 282 and 284 create another current mirror thatcopies the current in node 278 into node 286, the output node. Diodeconnected N-Channel MOS transistor 288 converts the current back into avoltage with a square root transfer function or a compressivenon-linearity. This is chosen to accentuate the low level currents andhence is suited to process a light touch at the sensor.

The increased sensitivity of the touch sensor system of the presentinvention allows for a lighter input finger touch which makes it easyfor human use. Increased sensitivity also makes it easier to use otherinput objects, like pen styli, etc. Additionally this sensitivity allowsfor a trade-off against a thicker protective layer, or differentmaterials, which both allow for lower manufacturing costs.

Greater noise rejection allows for greater flexibility in use andreduced sensitivity to spurious noise problems. Two techniques areemployed which allow derivation of the most noise-rejection benefit.

Due to the drive and sense techniques employed in the present invention,the data acquisition rate has been increased by about a factor of 30over the prior art. This offers several obvious side effects. First, forthe same level of signal processing, the circuitry can be turned offmost of the time and reduce power consumption by roughly a factor of 30in the analog section of the design. Second, since more data isavailable, more signal processing, such as filtering, and gesturerecognition, can be performed.

The sensor electronic circuit employed in the present invention is veryrobust and calibrates out process and systematic errors. It will processthe capacitive information from the sensor and provide digitalinformation to an external device, for example, a microprocessor.

Because of the unique physical features of the present invention, thereare several ergonomically interesting applications that were notpreviously possible. Presently a Mouse or Trackball is not physicallyconvenient to use on portable computers. The present invention providesa very convenient and easy-to-use cursor position solution that replacesthose devices.

In mouse-type applications, the sensor of the present invention may beplaced in a convenient location, e.g., below the "space bar" key in aportable computer. When placed in this location, the thumb of the usermay be used as the position pointer on the sensor to control the cursorposition on the computer screen. The cursor may then be moved withoutthe need for the user's fingers to leave the keyboard. Ergonomically,this is similar to the concept of the Macintosh Power Book with it'strackball, however the present invention provides a significantadvantage in size over the track ball. Extensions of this basic idea arepossible in that two sensors could be placed below the "space bar" keyfor even more feature control.

The computer display with it's cursor feedback is one small example of avery general area of application where a display could be a field oflights or LED's, a LCD display, or a CRT. Examples include touchcontrols on laboratory equipment where present equipment uses aknob/button/touch screen combination. Because of the articulatingability of this interface, one or more of those inputs could be combinedinto one of our inputs.

Consumer Electronic Equipment (stereos, graphic equalizers, mixers)applications often utilize significant front panel surface area forslide potentiometers because variable control is needed. The presentinvention can provide such control in one small touch pad location. AsElectronic Home Systems become more common, denser and more powerfulhuman interface is needed. The sensor technology of the presentinvention permits a very dense control panel. Hand Held TV/VCR/Stereocontrols could be ergonomically formed and allow for more powerfulfeatures if this sensor technology is used.

The sensor of the present invention can be conformed to any surface andcan be made to detect multiple touching points, making possible a morepowerful joystick. The unique pressure detection ability of the sensortechnology of the present invention is also key to this application.Computer games, "remote" controls (hobby electronics, planes), andmachine tool controls are a few examples of applications which wouldbenefit from the sensor technology of the present invention.

Musical keyboards (synthesizers, electric pianos) require velocitysensitive keys which can be provided by the pressure sensing ability ofthis sensor. There are also pitch bending controls, and other slideswitches that could be replaced with this technology. An even moreunique application comprises a musical instrument that creates notes asa function of the position and pressure of the hands and fingers in avery articulate 3-d interface.

The sensor technology of the present invention can best detect anyconducting material pressing against it. By adding a conductive foammaterial on top of the sensor the sensor of the present invention mayalso indirectly detect pressure from any object being handled,regardless of its electrical conductivity.

Because of the amount of information available from this sensor it willserve very well as an input device to virtual reality machines. It iseasy to envision a construction that allows position-monitoring in threedimensions and some degree of response (pressure) to actions.

While embodiments and applications of this invention have been shown anddescribed, it would be apparent to those skilled in the art that manymore modifications than mentioned above are possible without departingfrom the inventive concepts herein. The invention, therefore, is not tobe restricted except in the spirit of the appended claims.

What is claimed is:
 1. An object proximity sensor, including:atouch-sensitive transducer disposed on a substrate, said touch sensitivetransducer including a matrix of row conductive lines disposed in afirst direction and column conductive lines disposed in a seconddirection generally perpendicular to said first direction, said rowconductive lines and said column conductive lines insulated from oneanother, and an insulating layer disposed over said row conductive linesand said column conductive lines, said insulating layer forming a touchsurface, said insulating layer having a thickness selected to promotecapacitive coupling between a finger placed proximate to the touchsurface of said insulating layer and said row conductive lines and saidcolumn conductive lines; means for simultaneously injecting electricalcharge onto each of said row conductive lines, and for sensing arow-sense voltage created on each of said row conductive lines by saidelectrical charge onto each of said row conductive lines; means forsimultaneously injecting electrical charge onto each of said columnconductive lines, and for sensing a column-sense voltage created on eachof said column conductive lines by said electrical charge onto each ofsaid column conductive lines; and means for producing a set ofobject-sensed electrical signals related to said row-sense voltage andsaid column-sense voltage; and means for processing said set of rowelectrical signals and said set of column electrical signals to create aproximity electrical signal proportional to the proximity of said objectto said touch surface.
 2. The object proximity sensor of claim 1 whereinsaid row conductive lines are disposed on a first face of said substrateand said column conductive lines are disposed on a second face of saidsubstrate opposite said first face, said touch sensitive transducerfurther including a plurality of spaced-apart conductive sensor padsdisposed in a row and column matrix pattern on said substrate, each ofsaid sensor pads connected to a corresponding one of said row conductivelines or column conductive lines.
 3. The object proximity sensor ofclaim 1, further including:means for sensing a minimum no-objectproximate capacitance from among said row conductive lines, for sensinga minimum no-object proximate capacitance from among said columnconductive lines, for producing a set of minimum background electricalsignals related thereto; and means for subtracting said set of minimumbackground electrical signals from said set of object-sensed electricalsignals.
 4. The object proximity sensor of claim 1, furtherincluding:means for producing a set of average no-object-proximateelectrical signals related to an average no-object-proximate capacitancefrom among said row conductive lines and an average no-object-proximatecapacitance from among said column conductive lines; and means forsubtracting said set of average no-object-proximate electrical signalsfrom said set of object-sensed electrical signals.
 5. The objectproximity sensor of claim 1 wherein the ones of said sensor padsassociated with odd numbered ones of said row conductive lines aredisposed along a first set of column positions and the ones of saidsensor pads associated with even numbered ones of said row conductivelines are disposed at a second set of column positions offset from saidfirst set of column positions wherein said sensor pads form a closelypacked repetitive pattern wherein each pad is not in contact withadjoining pads.
 6. A method for providing an electrical signalrepresentative of the position of an object in a two dimensional sensingplane and of the proximity of the object to the two dimensional sensingplane, including the steps of:providing a sensing plane including amatrix of conductors arranged as a plurality of rows and columns ofspaced apart row conductive lines and column conductive lines, saidsensing plane having an inherent capacitance on the various ones of saidrow conductive lines and column conductive lines, said capacitancevarying with the proximity of an object to said row and columnconductors; simultaneously generating from among said row conductivelines a first electrical signal proportional to a no-object-proximatevalue of said capacitance when no object is proximate to said sensingplane from among said row conductive lines; simultaneously generatingfrom among said row conductive lines a corresponding second electricalsignal proportional to the value of said capacitance when an object islocated proximate to but not necessarily in contact with said sensingplane; subtracting each of said corresponding first electrical signalsfrom said second electrical signals to produce a set of row electricalsignals; simultaneously generating from among said column conductivelines a third electrical signal proportional to the no-object-proximatevalue of said capacitance when no object is proximate to said sensingplane from among said column conductive lines; simultaneously generatingfor each conductor in the column dimensions a corresponding fourthelectrical signal proportional to the value of said capacitance when anobject is located proximate to but not necessarily in contact with saidsensing plane; subtracting each of said corresponding third electricalsignals from said fourth electrical signals to produce a set of columnelectrical signals; encoding said set of row electrical signals and saidset of column electrical signals into electrical signals indicating theposition of said object in said row dimension and said column dimension;and processing said set of row electrical signals and said set of columnelectrical signals to create proximity electrical signal proportional toproximity of said object to said sensing plane.
 7. The method of claim 6wherein the step of encoding said set of row electrical signals and saidset of column electrical signals into electrical signals indicating theposition of said object in said row dimension and said column dimensioncomprises separately encoding said set of row electrical signals into afirst digital signal and encoding said set of column electrical signalsinto a second digital signal.
 8. The method of claim 6 wherein the stepof encoding said set of row electrical signals and said set of columnelectrical signals into electrical signals indicating the position ofsaid object in said row dimension and said column dimension comprisesseparately encoding said set of row electrical signals into a firstdigital signal and encoding said set of column electrical signals into asecond digital signal, and further including the step of encoding saidproximity electrical signal into a third digital signal.
 9. The methodof claim 6 further including the step of providing a signal when saidrow electrical signal for any of said row or column electrical signalexceeds a threshold value.
 10. The method of claim 6, wherein said firstand third electrical signals are proportional to the minimumno-object-proximate value of said capacitance from among said rowconductive lines and column conductive lines.
 11. The method of claim 6,wherein said first and third electrical signals are proportional to theaverage no-object-proximate value of said capacitance from among saidrow conductive lines and column conductive lines.